Self-calibrated power amplifier linearizers

ABSTRACT

An amplifier linearizer includes a signal adjuster having an internal signal, and an adaptation controller for monitoring the signal adjuster. The internal signal at an input to the adaptation controller is deemed a monitor signal. The adaptation controller generates a control signal for the signal adjuster by accounting for a difference between the internal and monitor signals.

CROSS-REFERENCE TO RELATED APPLICATIONS

[0001] This application claims priority to U.S. Patent Application No.60/301,978, filed Jun. 28, 2001.

FIELD OF THE INVENTION

[0002] This application generally pertains to, but is not limited to,linearizers used in power amplifiers, for example, RF power amplifiersused in wireless communication systems.

BACKGROUND OF THE INVENTION

[0003] RF power amplifiers, like most amplifiers, are substantiallylinear at small signal amplitudes. However, it is preferable to drivepower amplifiers near saturation to deliver significant output power ata reasonable efficiency. As the operation of a power amplifierapproaches saturation, it will become more nonlinear, and thus, exhibitmore distortion in its output. Consequently, numerous “linearizer”circuits have been developed over the years in an attempt to remove thepower amplifier's nonlinearity and thereby reduce the distortion in itsoutput. Because the characteristics of the power amplifier may changeover time and frequency, these linearizer circuits may be designed toadapt to present amplifier conditions. A generic power amplifierlinearizer is shown in FIG. 1, and uses either predistortion circuitry,feedforward circuitry, or a combination of both, to correct for thepower amplifier's nonlinearity. (The inclusion of this genericlinearizer in this Background Section is not intended to imply that thecircuit configuration shown therein, and variations thereof, are in theprior art.)

[0004] For example, a linearizer may use only a predistortion adjustercircuit p. As will be appreciated by those skilled in the art, in thislinearizer the signal adjuster circuit s is merely a delay line ideallymatching the total delay of the adjuster circuit p and the poweramplifier. In this case, the distortion cancellation circuit, comprisingthe distortion adjuster circuit d, the error amplifier and the delaycircuit, is not used—the output of the linearizer is the output of thesignal power amplifier. The goal of the adjuster circuit p is topredistort the power amplifier input signal so that the power amplifieroutput signal is proportional to the input signal of the linearizer.That is, the predistorter acts as a filter having a transfercharacteristic which is the inverse of that of the power amplifier,except for a complex constant (i.e., a constant gain and phase). Becauseof their serial configuration, the resultant transfer characteristic ofthe predistorter and the power amplifier is, ideally, a constant gainand phase that depends on neither frequency nor signal level.Consequently, the output signal will be the input signal amplified bythe constant gain and out of phase by a constant amount, that is,linear. Therefore, to implement such predistortion linearizers, thetransfer characteristic of the power amplifier is computed and apredistortion filter having the inverse of that transfer characteristicis constructed. Preferably, the predistortion filter should alsocompensate for changes in the transfer function of the power amplifier,such as those caused by degraded power amplifier components.

[0005] Other linearizers use feedforward circuitry to correct for thenonlinearity in the power amplifier. A feedforward linearizer usuallyuses a combination of signal adjuster circuit s 110 and distortionadjuster circuit d 111 as configured in FIG. 1 (in this linearizer,predistortion adjuster p 109 is not used). In an alternativeconfiguration, the signal adjuster circuit may be placed before thepower amplifier, i.e., as adjuster circuit p 109, an example of which isshown in FIG. 7. This latter configuration advantageously compensatesfor any additional signal distortion caused by the signal adjustercircuit, since it will be superimposed upon the distortion caused by thepower amplifier and be removed by a distortion cancellation circuit.Further, adjusters p 109, s 110, and d 111 may be used simultaneously tolinearize the power amplifier.

[0006] As shown in FIG. 1, a feedforward linearizer comprises two maincircuits: a signal cancellation circuit 101 and a distortioncancellation circuit 102. The RF signal is input to the signal poweramplifier 103, which as discussed above, is assumed to be operating in anon-linear range and thus distorting the output signal. The signalcancellation circuit 101 ideally subtracts a linear estimate of the RFsignal from the distorted power amplifier output signal so that only thenonlinear distortion signal (or “error signal”) (v_(e)) remains. As willbe appreciated to those skilled in the art, the signal pickoff points,the adder 104, and the subtractors 106 and 107 shown in FIG. 1 and otherfigures may be implemented by directional couplers, splitters orcombiners, as appropriate. In the distortion cancellation circuit 102,the distortion signal is adjusted and amplified by error amplifier 108to match the distortion signal component of the power amplifier outputsignal delayed by delay 112. The amplified distortion signal is thensubtracted from the output of delay 112 by subtractor 107 to provide thelinearizer output signal v_(o). The linearizer output signal is asubstantially distortion-free amplified RF signal, the output that wouldhave been obtained if the power amplifier were truly linear.

[0007] Generally, the adjuster circuits discussed above do notnecessarily all have the same structure—adjuster circuits p 109, s 110and d 111 may all be implemented with different circuitry. For example,the adjuster circuit p 109 may be a nonlinear polynomial filter, whilethe adjuster circuits s 110 and d 111 may be finite impulse response(FIR) filters. In addition, some methods of controlling these adjustercircuits may employ pilot (tone) signals generated by an optional pilotsignal generator 113.

[0008] The relationship of the input and output signals of an adjustercircuit depends on the settings of one or more parameters of thatadjuster circuit, as will be discussed in further detail below. Duringadaptation, the values of one or more internal signals of an adjustercircuit are used to determine appropriate settings for its parameters.As shown in FIG. 1, an “adaptation controller” 114 monitors the errorand output signals v_(e) and v_(o), and in some cases, the internaladjuster signals. (In FIG. 1 and other figures, a stroke on an arrowdenotes a multiplicity of signals or a multiplicity of parameters, asthe case may be.) On the basis of the monitored signal values, and inaccordance with the adaptation algorithm, the adaptation controller setsthe adjuster circuit parameters.

[0009] For example, a three-branch adaptive polynomial predistortionadjuster circuit p 109 is shown in FIG. 2. The upper branch 200 islinear, while the middle branch has a nonlinear cubic polynomial filter201 and the lower branch has a nonlinear quintic polynomial filter 202,the implementation of which nonlinear filters is well known to thoseskilled in the art. Each branch also has a complex gain adjuster(“CGA”), respectively 203, 204, and 205, to adjust the amplitude andphase of the signal as it passes therethrough. By setting the parameters(GA, GB) of each of the CGAs, a polynomial relationship between theinput and output of the adjuster circuit can be established tocompensate for a memoryless nonlinearity in the power amplifier. Theadaptation controller, via a known adaptation algorithm, uses the inputsignal, the output of the nonlinear cubic polynomial filter, the outputof the nonlinear quintic polynomial filter, and the error signal (thepower amplifier output signal minus an appropriately delayed version ofinput signal) to generate the parameters (GA, GB) for the three CGAs.Generally, the adaptation algorithm is selected to minimize a certainparameter related to the error signal (for example, its power over apredetermined time interval). Examples of such adaptation algorithms aredescribed in more detail below.

[0010] Two possible CGA implementations are respectively shown in FIGS.3A and 3B. The implementation shown in FIG. 3A uses polar controlparameters GA and GB, where GA sets the amplitude of the attenuator 301,while GB sets the phase of the phase shifter 302. The implementationshown in FIG. 3B uses Cartesian control parameters, also designated GAand GB, where GA sets the real part of the complex gain, while GB setsthe imaginary part of the complex gain. In this implementation, theinput signal I is split into two signals by splitter 306, one of whichis then phase-shifted by 90 degrees by phase shifter 303, while theother is not. After GA and GB are applied by mixers or attenuators 305and 304 respectively, the signals are summed by combiner 307 to producethe CGA output signal O. U.S. Pat. No. 6,208,207 describes, in part, thelinearization of these mixers and attenuators, so that desired values ofcomplex gain can be predictably obtained by appropriate setting of thecontrol voltages GA and GB.

[0011] For example, a three-branch adjuster circuit s 110 and athree-branch adjuster circuit d 111 in a feedforward linearizer areshown in FIG. 4. Feedforward linearizers having one or more branches inthe adjuster circuits are described in U.S. Pat. Nos. 5,489,875 and6,208,207, both of which are incorporated herein by reference. Eachbranch within the circuits 110 and 111, labeled “FIR adjuster”, includesa delay element (i.e., delays 401, 403 and 405 in the adjuster s 110,and delays 407, 409, and 411 in adjuster d 111) and a CGA (i.e., CGAs402, 404 and 406 in the adjuster s 110, and CGAs 408, 410, and 412 inadjuster d 111). The delays in each branch may be different, and the sumof the parallel branches act as an analog FIR filter (also known as ananalog transversal filter).

[0012] Appropriate settings of the parameters (GA, GB) of the CGAs allowthe first FIR adjuster circuit 110 to mimic the linear portion of thepower amplifier response, including the effects of amplifier delay andother filtering, and for frequency dependence of its own components.Ideally, the amplifier nonlinear distortion is revealed at the output ofthe subtractor following the first FIR adjuster circuit (v_(e)).Appropriate settings of the parameters (GA, GB) of the second FIRadjuster circuit 111 allow it to compensate for delay and otherfiltering effects in the amplifier output path and for frequencydependence in its own components, and to subtract a replica of thenonlinear distortion from the delayed amplifier output. The adaptationcontroller 114 of FIG. 4, via a known adaptation algorithm, uses theinternal signals of the branches of the signal and distortion adjustercircuits s 110 and d 111 and their respective error signals v_(e) andv_(o), to compute GA and GB for each of the CGAs in the signal anddistortion adjuster circuits 110 and 111. In this fashion, thelinearization circuit compensates for an amplifier nonlinearity withmemory. Examples of such adaptation controllers can be found in U.S.Pat. Nos. 5,489,875 and 6,208,207.

[0013] The linearizer circuits of the prior art, however, ignore aphenomenon that often determines the success or failure of theadaptation controller—the monitored signals, as measured by theadaptation controller, are not necessarily equal to their counterpartinternal signals within the adjuster circuits, or to the actual errorand output signals v_(e) and v_(o), as the case may be. The reason isthat the cables, circuit board traces, and other components in thesignal paths that convey the internal adjuster circuit signals, or theerror and output signals v_(e) and v_(o), to the adaptation controllerintroduce inadvertent phase and amplitude changes into those signals.The true situation is represented in FIGS. 6 and 8, where these phaseand amplitude changes are generically modeled as “observation filters”(601-603, 804, 805). In addition, H_(p)(f) 601, H_(s)(f) 602 andH_(d)(f) 603 may each be considered to comprise a bank of “observationsubfilters,” such as h_(p1)(f), h_(p2)(f), etc., each observationsubfilter modeling the transformation of a particular internal signal ofan adjuster circuit into a corresponding monitor signal (for example,one observation subfilter per branch of a multibranch adjuster). Thecharacteristics of these observation filters or subfilters are initiallyunknown.

[0014] In the simplest case, the observation filters or subfilters mayrepresent fixed amplitude and phase changes on each of the signal paths.In a more complex case, however, the amplitude and phase changes, andthus the observation filters, can be frequency-dependent. For example, athree-branch signal adjuster p 109 located in front of the poweramplifier 103 is shown in FIG. 7. This adjuster circuit is constructedso that each branch k (k=0, 1, 2) contains a frequency-dependent filterg_(k)(f) (701, 703 or 705), which serves as a generalization of thedelay elements of an FIR adjuster, and a CGA (702, 704 or 706). (Themention of these general branch filters g_(k)(f) in this BackgroundSection is not intended to imply that their use in FIR adjuster circuitsis known in the prior art; rather, such use is intended to be within thescope of the present invention.) Each observation subfilter h_(k)(f)(707, 708 or 709) of observation filter H_(p)(f) 601 models thetransformation of the internal signal on branch k into the correspondingmonitor signal used by the adaptation controller.

[0015] It should be understood that placement of filters (or filterbanks) as shown in FIG. 6 is just one of many ways to model thedifference (inequality) between the internal signals and the monitorsignals. Nevertheless, the observation filters shown in FIG. 6 aresufficient, because other ways of modeling the difference betweeninternal signals and their corresponding monitor signals are equivalentto the representation thereto. For example, FIG. 8 shows a linearizercircuit that includes an observation filter h_(em)(f) 804 in the path ofthe error signal v_(e) output from the first subtractor 106, and anobservation filter h_(om)(f) 805 in the path of the RF output signalv_(o), output from the second subtractor 107, to the adaptationcontroller 114. These observation filters can be transformed to therepresentation shown in FIG. 6 by including the effect of h_(em)(f) 804in the branch paths of adjuster circuits p 109 and s 110, and ofh_(om)(f) 805 into the branch path of adjuster circuit d 111 and thedistortion cancellation circuit reference branch.

[0016] The severity of the problem caused by the differences between theinternal adjuster signals and their corresponding monitor signals can beillustrated by a simple example. FIG. 9 illustrates the signalcancellation circuit 101 of a single-branch feedforward linearizer.Specifically, the signal adjuster circuit s 110 includes a single delay901 followed by a CGA 902. The adaptation controller 114 uses a known“stochastic gradient” algorithm (see, for example, the gradientadaptation controller disclosed in U.S. Pat. No. 5,489,875) to correlateusing bandpass correlator 903 the error signal at the output of thesubtractor with the monitored replica of the internal signal of theadjuster circuit, both of which are bandpass signals. The controllerintegrates the result using integrator 905, via loop gain amplifier 904,to provide CGA parameters GA and GB. The internal structure of a knownbandpass correlator is depicted in FIG. 10, and includes a phase shifter1001, mixers 1002 and 1003, and bandpass filters (or integrators) 1004and 1005 (for a description of the operation of such a bandpasscorrelator, see FIG. 3 of U.S. Pat. No. 5,489,875 and the textcorresponding thereto). In the idealized situation considered in theprior art, the monitor signal and the internal signal are equal, withH_(s)(f)=h=1 (906) at all frequencies, and the correlation result is astochastic estimate of the gradient of the error signal power withrespect to the CGA parameters. That is, the correlation result isproportional to the change in CGA parameter settings that would resultin the greatest increase in error signal power. Sign reversal andintegration causes the CGA parameters to be corrected in the directionthat most decreases the error signal power, and the adaptation loopconverges correctly to the setting of GA and GB that minimizes errorsignal power, with a time constant determined by the value K of the loopgain amplifier 904.

[0017] To continue this example, when the linearizer is implemented, itwill differ from the ideal case in that the monitor signal (bandpasssignal 1 of FIG. 10) will likely have unknown phase and amplitude shiftswith respect to the internal adjuster signal. These shifts arerepresented by the complex variable h (906) in FIG. 9. If h has a phaseshift of 180 degrees, then the correlation result will be negated, andthe correction to the CGA parameters will be made in a direction thatmaximally increases, rather than decreases, the error signal power.This, in turn, will cause the circuit to diverge from its ideal setting.More generally, a phase shift in h of over 90 degrees will cause thecircuit to diverge. A phase shift value of greater than zero degrees,but less than 90 degrees, will allow the circuit to converge to itsideal setting, but with decreasing rapidity as it approaches 90 degrees.

[0018] The same problem may afflict adaptation controllers based onother algorithms that exploit the relationships among monitored signalsin order to make corrections to CGA settings. For example, a leastsquares (“LS”) algorithm or a recursive LS algorithm may also diverge(or converge more slowly) under the same phase shift conditions as setforth above for the stochastic gradient algorithm.

[0019] In a multibranch adjuster circuit, for example, the polynomialpredistorter circuit of FIG. 2 or the feedforward circuit of FIGS. 4 and6, there are further consequences of the lack of equality between aninternal adjuster signal and its corresponding monitor signal. Signalscarried on the branches of an adjuster tend to be highly correlated,making stochastic gradient adaptation slow. The remedy is lineartransformation of the multiple branch signals to produce a multiplicityof decorrelated signals. The decorrelated signals, or modes, are thenadapted individually to provide a much faster convergence. However, thelack of equality between internal and corresponding monitor signals, asmodeled by the filter banks in FIG. 6, reduces the ability todecorrelate those signals completely. This results in branch signalswith residual correlation, thus reducing the benefits of decorrelation.Furthermore, if the internal and monitor signals are not equal,unfavorable phase and amplitude relationships among the filters maycause one or more of the decorrelation mode adaptations to diverge,preventing adaptation altogether.

[0020] In addition, in a stochastic gradient controller, if wide powerdisparities exist among the decorrelated signals, stronger signals mayinterfere or “mask” the weaker signals, degrading the latter and slowingadaptation. To reduce this masking problem, a “partial gradient”algorithm may be used by the adaptation controller (see, for example,the partial gradient adaptation controller disclosed in U.S. Pat. No.5,489,875), in which the correlation between two bandpass signals isapproximated as a sum of partial correlations taken over limitedbandwidths at selected frequencies. By making the frequenciesselectable, correlations may be calculated at frequencies that do notcontain strong signals, so that the strong signals do not mask the weaksignals. In addition, a digital signal processor (DSP) may be used toperform correlation, because the correlations are taken over limitedbandwidths. This eliminates the DC offset that otherwise appears in theoutput of a correlator implemented by directly mixing two bandpasssignals.

[0021]FIG. 11 illustrates a partial correlator, in which localoscillators 1101 and 1102 select the frequency of the partialcorrelation. Frequency shifting and bandpass filtering are performed bythe mixer/bandpass filter combinations 1103/1107, 1104/1108, 1105/1109,and 1106/1110. The signals output by the bandpass filters 1109 and 1110are digitally converted, respectively, by A/D converters 1111 and 1112.Those digital signals are bandpass correlated by DSP 1113 to produce thereal and imaginary components of the partial correlation. (See, forexample, FIG. 9 of U.S. Pat. No. 5,489,875 for a description of theoperation of a partial correlator similar to that shown in FIG. 11herein.) However, as in the case of the stochastic gradient adaptationcontroller, a lack of equality between the internal signals and theircorresponding monitor signals (for example, bandpass signal 1) may causeeither divergence or slowed convergence of the partial gradientadaptation controller of FIG. 11.

[0022] Accordingly, self-calibrated power amplifier linearizers aredesired to compensate for the lack of equality between the internaladjuster signals and their corresponding monitor signals, and toovercome the resulting divergence, or slowed convergence, of theadaptation controllers used therein.

SUMMARY OF THE INVENTION

[0023] In one aspect of the presented invention, an amplifier linearizerincludes a signal adjuster having an internal signal, and an adaptationcontroller for monitoring the signal adjuster. The internal signal at aninput to the adaptation controller is deemed a monitor signal. Theadaptation controller generates a control signal for the signal adjusterby accounting for a difference between the internal and monitor signals.

[0024] This and other aspects of the invention may be ascertained fromthe detailed description of the preferred embodiments set forth below,taken in conjunction with the one or more of the following drawings.

BRIEF DESCRIPTION OF DRAWINGS

[0025]FIG. 1 is a block diagram of a generic power amplifier linearizercircuit.

[0026]FIG. 2 shows an example of a polynomial predistortion linearizercircuit.

[0027]FIGS. 3A and 3B respectively show two configurations of a complexgain adjuster.

[0028]FIG. 4 shows an example of a multibranch feedforward linearizercircuit.

[0029]FIG. 5 shows a hybrid predistorter and feedforward linearizercircuit.

[0030]FIG. 6 is a block diagram of a generic power amplifier linearizercircuit, in which observation filters are included to model the lack ofequality between the internal adjuster signals and the correspondingmonitor signals used by the adaptation controller.

[0031]FIG. 7 is a generic FIR signal adjuster circuit showing a bank ofobservation subfilters to model the lack of equality between theinternal signal adjuster circuit signals and the corresponding monitorsignals used by the adaptation controller.

[0032]FIG. 8 is a block diagram of a generic power amplifier linearizercircuit, in which observation filters are included to model the lack ofequality between the internal adjuster signals, error output signalv_(e) and RF output signal v_(o), and the corresponding monitor signalsused by the adaptation controller.

[0033]FIG. 9 is an example of a one-branch feedforward signal adjustercircuit, and a stochastic gradient adaptation controller using abandpass correlator, in which observation filter h is included to modelthe lack of equality between the internal signal adjuster signal and thecorresponding monitor signal used by the adaptation controller.

[0034]FIG. 10 is an example of the bandpass correlator used by theadaptation controller shown in FIG. 9.

[0035]FIG. 11 is an example of a partial bandpass correlator used in apartial gradient adaptation controller.

[0036]FIG. 12 is a two-branch feedforward circuit with observationfilters included in the monitor signal paths of the FIR adjusterbranches.

[0037]FIG. 13 is a two-branch feedforward signal cancellation circuitwith observation filters for the adjuster branches.

[0038]FIG. 14 is a multibranch predistorter containing generalnonlinearities with frequency dependence.

DETAILED DESCRIPTION OF THE PREFERRED EMBODIMENTS

[0039] The embodiments of the present invention are directed toself-calibration techniques by which the adaptation controller (1)determines the frequency responses of the observation filters betweenthe internal signals and the monitor signals (and hence, the frequencyresponses of the underlying circuit components modeled by thoseobservation filters) and (2) corrects for these responses, so that themonitored signals are substantially representative of the internalsignals of the adjuster circuits. The benefits of self-calibration are(1) reliable adaptation, without risk of divergence; (2) fasteradaptation than without self-calibration; and (3) access to methods thatdecorrelate the internal signals, explicitly or implicitly, to allowfast adaptation of all the signal modes. This self-calibration can beperformed upon the initialization of the linearizer (for example, atdevice turn-on), or from time-to-time as needed. Further, theself-calibration is performed by the linearizer itself without humanintervention.

[0040]FIG. 12 illustrates a feedforward linearizer in which an FIRadjuster s 1210 in the signal cancellation circuit is located parallelto the power amplifier 103 (the case where the FIR adjuster s is inseries with the power amplifier is discussed further below). The FIRadjuster s has two branches, the upper including a delay 1230 and a CGA1231, and the lower including a delay 1232 and a CGA 1233. The outputsof the CGAs 1231 and 1233 are summed by the combiner 1238. Thefeedforward linearizer also has an FIR adjuster d 1211 in the distortioncancellation circuit. The FIR adjuster d also has two branches, theupper including a delay 1234 and a CGA 1235, and the lower including adelay 1236 and a CGA 1237. The outputs of the CGAs 1235 and 1237 aresummed by the combiner 1239. Although FIG. 12 illustrates FIR adjusterswith two branches, the self-calibration techniques of the presentinvention apply to adjusters with one or more branches. In addition,these self-calibration techniques are described below with respect toFIR adjuster circuits for sake of clarity. However, those skilled in theart will recognize that these techniques apply to a wide range ofadjuster circuits having structures as described above, including, butwithout limitation, FIR adjusters, polynomial adjusters, and adjusterswith general filters on the branches.

[0041] In the circuit shown in FIG. 12, the complex gains (amplitude andphase) of the observation filters h_(s0)(f) 1220, h_(s1)(f) 1221,h_(e0)(f) 1222 and h_(e1)(f) 1223 are first determined. Then those gainsare used to adjust the respective monitor signals v_(am0), v_(am1),v_(bm0) and v_(bm1) so they are representative of the correspondinginternal adjuster signals v_(a0), v_(a1), v_(b0) and v_(b1).

[0042] In one embodiment, it is assumed that the observation filters donot depend on frequency, so that they are each characterized by a singlecomplex gain, i.e., h_(s0), h_(s1), h_(e0), and h_(e1). With referenceto FIG. 12, the adaptation controller 1214 determines the complex gainh_(s0) as follows:

[0043] (1) set the power amplifier 103 in standby mode, so that itsoutput is zero;

[0044] (2) set the CGA 1233 complex gain a₁ to zero through anappropriate choice of the control voltages, so that the correspondingCGA output is zero;

[0045] (3) set the CGA 1231 complex gain a₀ to some nominal value a₀′through appropriate choice of control voltages;

[0046] (4) apply an input signal with components at frequency f₁ to theamplifier, or use an internal pilot signal generator 113 to generate atone for calibration;

[0047] (5) in the adaptation controller, use a bandpass correlator (forexample, the bandpass correlator shown in FIG. 10) to produce thecorrelation of signal v_(e) with monitor signal v_(am0); the result is:

[0048] C_(eam0)=a₀′·h_(s0) ^(*)·P_(a0), where the asterisk denotescomplex conjugation and P_(a0) denotes the power of signal v_(a0);

[0049] (6) in the adaptation controller, use a bandpass correlator toproduce the correlation of monitor signal v_(am0) with itself, theresult is:

[0050] C_(am0)=|h_(s0)|²·P_(a0), where the bars denote the magnitude ofa complex quantity; and

[0051] (7) determine the observation filter gain as:

h _(s0) =a ₀ ′·C _(am0) /C _(eam0).

[0052] gain h_(s1) is determined in a similar fashion with branch “0”set to zero (a₀=0) and branch “1” enabled (a₁ =a₁′).

[0053] The adaptation controller 1214 determines the complex gain h_(e0)as follows:

[0054] (1) set the power amplifier in standby mode, so that its outputis zero, and set at least one of the CGA gains (a₀ or a₁) in the signalFIR adjuster s to a non-zero value, so that the power of the errorsignal P_(e) is non-zero;

[0055] (2) set the CGA 1237 complex gain b₁ to zero through anappropriate choice of the control voltages, so that the correspondingCGA output is zero;

[0056] (3) set the CGA 1235 complex gain b₀ to some nominal value b₀′through appropriate choice of control voltages;

[0057] (4) apply an input signal with components at frequency f₁ to theamplifier, or use an internal pilot signal generator 113 to generate atone for calibration;

[0058] (5) in the adaptation controller, use a bandpass correlator toproduce the correlation of signal v_(o) with monitor signal v_(bm0); theresult is:

[0059] C_(obm0)=b₀′·h_(e0) ^(*)·P_(b0), where the asterisk denotescomplex conjugation and P_(b0) denotes the power of signal v_(b0);

[0060] (6) in the adaptation controller, use a bandpass correlator toproduce the correlation of monitor signal v_(bm0) with itself; theresult is:

[0061] C_(bm0)=|h_(e0)|²·P_(b0), where the bars denote the magnitude ofa complex quantity; and

[0062] (7) determine the observation filter gain as:

h _(e0) =b ₀ ′·C _(bm0) /C _(obm0).

[0063] The gain h_(e1) is determined in a similar fashion with branch“0” set to zero (b₀=0) and branch “1” enabled (b₁=b₁′).

[0064] In another embodiment, it is assumed that the observation filtersdepend on frequency. Consequently, to approximate their frequencyresponses, the adaptation controller determines their gains at aselected set of N frequencies f_(i), i=1, 2, . . . , N. The adaptationcontroller 1214 determines the gain h_(s0)(f₁) at frequency f₁ asfollows:

[0065] (1) set the power amplifier 103 to standby mode, so that itsoutput is zero;

[0066] (2) set the CGA 1233 gain of a₁ to zero through appropriatechoice of the control voltages, so that the CGA output is zero;

[0067] (3) set the CGA 1231 gain a₀ to some nominal value a₀′ throughappropriate choice of control voltages;

[0068] (4) apply an input signal with components at frequency f₁ to theamplifier, or use an internal pilot signal generator 113 set tofrequency f₁;

[0069] (5) use a partial correlator (for example, the partial correlatorshown in FIG. 11), with local oscillators set to select frequency f₁, toproduce the correlation of signal v_(e) with monitor signal v_(am0); theresult is:

[0070] C_(eam0)(f₁)=a₀·h_(s0) ^(*)(f₁)·P_(a0)(f₁), where P_(a0)(f₁)denotes the power of signal v_(a0) at frequency f₁;

[0071] (6) use a partial correlator, with local oscillators set toselect frequency f₁, to produce the correlation of monitor signalv_(am0) with itself; the result is:

C _(am0)(f ₁)=|h _(s0)(f ₁)|² ·P _(a0)(f ₁);

[0072] (7) determine the observation filter gain at frequency f₁ as:

h _(s0)(f ₁)=a ₀ ′C _(am0)(f ₁)/(C _(eam0)(f ₁))

[0073] Similarly, the adaptation controller 1214 determines the gainh_(e0)(f₁) at frequency f₁ as follows:

[0074] (1) set the power amplifier 103 to standby mode, so that itsoutput is zero, and set at least one of the CGA gains (a_(o), a₁) in thesignal cancellation circuit to a non-zero value, so that the power ofthe error signal P_(e)(f₁) is non-zero;

[0075] (2) set CGA 1237 gain b₁ to zero through appropriate choice ofthe control voltages, so that the corresponding CGA output is zero;

[0076] (3) set the CGA 1235 gain b₀ to some nominal value b₀′ throughappropriate choice of control voltages;

[0077] (4) apply an input signal with components at frequency f₁ to theamplifier, or use an internal pilot signal generator 113 set tofrequency f₁;

[0078] (5) use a partial correlator, with local oscillators set toselect frequency f₁, to produce the correlation of signal v_(o) withmonitor signal v_(bm0); the result is:

C _(obm0)(f _(i))=b₀ ′·h _(e0) ^(*)(f ₁)·P _(b0)(f ₁), where P _(b0)(f₁),

[0079] denotes the power of signal v_(b0) at frequency f₁;

[0080] (6) use a partial correlator, with local oscillators set toselect frequency f₁, to produce the correlation of monitor signalv_(bm0) with itself; the result is:

C _(bm0)(f _(j))=|h _(e0)(f ₁)² ·P _(b0)(f ₁);

[0081] (7) determine the observation filter gain at frequency f₁ as:

h _(e0)(f ₁)=b ₀ ′C _(bm0)(f ₁)/(C _(obm0)(f ₁)).

[0082] The complex gains h_(s0)(f₁) and h_(e0)(f_(i)) at frequenciesi=2, 3, . . . , N are determined similarly. The frequency responses ofthe remaining observation filters h_(s1)(f) and h_(e1)(f) are determinedby selecting them one at a time through choice of CGA gains, and thenrepeating the above-described methods for each frequency f_(i), i=1, 2,. . . , N.

[0083] In the above procedures, the frequencies f_(i) at whichcalibration is obtained are the same for the signal and distortioncancellation circuits. However, the frequencies f_(i) at whichcalibration is obtained may differ between the signal and distortioncancellation circuits.

[0084] To self-calibrate a single-branch adjuster circuit, step (2) ofthe above-described methods is eliminated. To self-calibrate an adjustercircuit having more than two branches, one branch at a time is enabled,while all others are set to zero, until all corresponding observationfilter gains are determined.

[0085] Computing the complex gains of the observation filters for thesignal cancellation circuit may be done without computing the same forthe distortion cancellation circuit, and vice-versa. Also, computing thecomplex gains of one or more of the observation filters for thedistortion cancellation circuit may be done prior to computing the samefor the signal cancellation circuit, and vice-versa.

[0086] Moreover, although the adjusters in this embodiment had delaylines in the branches, as shown in FIG. 12, the procedures forestimating the observation filter complex gains, whether they befrequency-independent or frequency-dependent, are equally applicable toadjusters constructed with filters of any type in place of one or moreof the delay lines 1230, 1232, 1234 and 1236.

[0087] Once the observation filter complex gains are computed, thecorresponding monitor signals need to be appropriately adjusted. Theadaptation controller divides the monitor signals by the respectiveobservation filter complex gains (either frequency-independent orfrequency-dependent, as the case may be) to approximate the trueinternal adjuster circuit signals. For example, for branch k andfrequency f_(i), the controller calculates:

v _(ak) =v _(amk) /h _(sk)(f ₁); and

v _(bk) =v _(bmk) /h _(ek)(f ₁).

[0088] Once these self-calibration procedures are performed, the effecton convergence is dramatic. This is particularly true when a partialcorrelator is implemented using a DSP. Convergence is reliable androbust in the face of amplitude and phase changes introduced by thecables, circuit board traces, and other components in the signal pathsthat convey the internal signals to the adaptation controller. Moregenerally, any adaptation algorithm, such as stochastic gradient, withor without decorrelation of the branch signals, or least squares, can bemade insensitive to these amplitude and phase changes, because theadaptation controller can always recover the internal signals from themonitor signals by dividing them by the corresponding determinedobservation filter complex gains.

[0089] Variations of the self-calibration procedures of the presentinvention will be evident to those skilled in the art. Two are listedhere for illustrative purposes.

[0090] In one variation (described here only for a two-branch signaladjuster circuit, but equally applicable to a two-branch distortionadjuster circuit, as appropriately modified in accord with the methodfor computing h_(e0)(f_(i)) and h_(e1)(f_(i)) described above), theadaptation controller 1214 simultaneously determines the frequencyresponses of all observation filters at frequency f_(i) as follows:

[0091] (1) set the power amplifier 103 to standby mode, so that itsoutput is zero;

[0092] (2) set all the CGA gains to non-zero nominal values a_(k)′, k=0. . . K−1, where k indexes the branch and there are K branches (in FIG.12, the circuit is illustrated for K=2 branches);

[0093] (3) apply an input signal with components at frequency f₁ to theamplifier, or use an internal pilot signal generator 113 set tofrequency f_(i);

[0094] (4) use partial correlators to measure all the pairwisecorrelations among the monitor signals at frequency f_(i); for K=2, thisresults in:

C _(am0)(f _(i))=P _(a0)(f _(i))·|h _(s0)(f _(i))|²

C _(am1)(f _(i))=P _(a1)(f _(i))·|h _(s1)(f _(i))|²

C _(am01)(f _(i))=P _(a01)(f _(i))·h _(s0)(f _(i))·h _(s1) ^(*)(f _(i))

[0095]  where P_(a0)(f_(i)) denotes the power of v_(a0) at f_(i),P_(a1)(f_(i)) denotes the power of v_(al) at f_(i), and P_(a01)(f_(i))denotes the “crosspower” (the correlation) of v_(a0) and v_(a1) atf_(i);

[0096] (5) use partial correlators to measure the correlation betweenthe error signal and each of the monitor signals; for K=2, this resultsin

C _(eam0)(f _(i))=a ₀ ′·P _(a0)(f _(i))·h _(s0) ^(*)(f _(i))+a ₁ ′·P_(a01)(f _(i))·h _(s0) ^(*)(f _(i))

C _(eam1)(f _(i))=a ₀ ′·P _(a01)(f _(i))·h _(s1) ^(*)(f _(i))+a ₁ ′·P_(a1)(f _(i))·h _(s1) ^(*)(f _(i))

[0097]  which becomes, after substitution for the powers, a set ofequations in the observation filter gains:

C _(eam0)(f _(i))=a ₀ ′·C _(am0)(f _(i))·h _(s0)(f _(i))⁻¹ +a ₁ ′·C_(a01)(f _(i))·h _(s1)(f _(i))⁻¹

[0098] (6) solve the set of equations for h_(s0)(f_(i))⁻¹ andh_(s1)(f_(i))⁻¹; take their reciprocals to obtain the desired frequencyresponses h_(s0)(f_(i)) and h_(s1)(f_(i)).

[0099] The method extends in a straightforward way to signal ordistortion adjuster circuits with more than two branches.

[0100] In a second variation, described here only for the signaladjuster circuit (but equally applicable to distortion adjuster,appropriately modified as described above), the adaptation controller1214 determines the responses of the observation filters at frequency fiwithout putting the power amplifier 103 in standby mode. It will bedescribed here only for one of the branches (branch k):

[0101] (1) set all of the CGA gains to zero;

[0102] (2) apply an input signal with components at frequency f_(i) tothe amplifier, or use an internal pilot signal generator 113 set tofrequency f_(i); the power of the signal is set to operate the poweramplifier at a nominal operating point;

[0103] (3) use a partial correlator, with local oscillators set toselect frequency f_(i), to produce the correlation of signal v_(e) withmonitor signal v_(amk); the result is a bias term C′_(eamk)(f_(i));

[0104] (4) set the CGA gain a_(k) to some nominal value a_(k)′ throughappropriate choice of control voltages;

[0105] (5) use a partial correlator, with local oscillators set toselect frequency f_(i), to produce the correlation of signal v_(e) withmonitor signal v_(amk); the result is

C _(eamk)(f _(i))=a _(k) ′h _(sk) ^(*)(f _(i))·P _(ak)(f _(i))+C′_(eamk)(f _(i));

[0106] (6) use a partial correlator, with local oscillators set toselect frequency f_(i), to produce the correlation of monitor signalv_(amk); the result is

C _(amk)(f ₁)=|h _(sk)(f _(i))|² ·P _(ak)(f _(i));

[0107] (7) determine the observation filter gain at frequency f_(i) as

h _(sk)(f _(i))=a _(k) ′·C _(amk)(f _(i))/(C _(eamk)(f _(i))−C′_(eamk)(f _(i))).

[0108] In another aspect of the present invention, the adjuster circuit1309 precedes the power amplifier 103, as shown in FIG. 13, an expandedversion of FIG. 7. The branch filters h_(c0)(f) 1330 and h_(c1)(f) 1332can be as simple as delays or as complex as general linear filters.These filters respectively precede CGAs 1331 and 1332, the outputs ofwhich are summed by combiner 1334. In this model, the amplifier gain isincluded in the branch filter responses. The filter h_(r)(f) 1310 in thereference branch may also be simple or complex; even if such a filter isnot inserted explicitly, h_(r)(f) 1310 represents the response of thebranch. The RF switch 1340 is optional; as explained below, its presenceor absence gives rise to two embodiments. The objective in both cases isto determine the responses of the observation filters h_(p0)(f) 1320 andh_(p1)(f) 1321 at selected frequencies.

[0109] In the first of such embodiments, the RF switch 1340 is absentand there is an unobstructed path from the output of filter h_(r)(f)1310 to the input of the subtractor 106. To determine the responseh_(pk)(f_(i)) of the observation filter k at frequency f_(i) theadaptation controller 1314 performs the following actions:

[0110] (1) set the power amplifier to standby mode, so that its outputis zero;

[0111] (2) apply an input signal containing the frequency components atfrequency f_(i) or use an internal pilot signal generator set tofrequency f_(i); the power of the signal is set to operate the poweramplifier at a nominal operating point;

[0112] (3) use a partial correlator, with local oscillators set toselect frequency f_(i), to produce the correlation of signal v_(e) withmonitor signal v_(cmk); the result is a bias term:

C′ _(ecmk)(f _(i))=−h_(r)(f _(i))·h _(pk) ^(*)(f ₁)·h_(ck) ^(*)(f_(i))·P _(in)(f _(i)), where

[0113]  P_(in)(f_(i)) is the input power at frequency f_(i);

[0114] (4) restore the power amplifier to operational mode;

[0115] (5) set the branch-k CGA gain to some nominal value c′_(k); setall other CGA gains to zero;

[0116] (6) use a partial correlator, with local oscillators set toselect frequency f_(i), to produce the correlation of signal v_(e) withmonitor signal v_(cmk); the result is:

C _(ecmk)(f _(i))=(c′ _(k) ·h _(ck)(f _(i))−h _(r)(f _(i)))·h _(pk)^(*)(f _(i))·h _(ck) ^(*)(f _(i))·P _(in)(f _(i));

[0117] (7) use a partial correlator, with local oscillators set toselect frequency f_(i), to produce the correlation of monitor signalv_(cmk) with itself; the result is:

C _(cmk)(f _(i))=|h _(pk)(f _(i))|² ·|h _(ck)(f _(i))|² ·P _(in)(f_(i));

[0118] (8) determine the branch-k observation filter response atfrequency f_(i) as:

h _(pk)(f _(i))C′ _(k) c _(cmk)(f _(i))/(C _(ecmk)(f _(i))−C′_(ecmk)(f_(i))).

[0119] The observation filter responses for the other filters aredetermined similarly. In the second embodiment of this aspect of thepresent invention, the RF switch 1340 is present. As will be seen, itsimplifies the calibration procedure significantly. To determine theresponse h_(pk)(f_(i)) of the observation filter k at frequency f_(i),the adaptation controller performs the following actions:

[0120] (1) open the RF switch 1340, thereby disconnecting the filterh_(r)(f) 1310 from the subtractor 106;

[0121] (2) apply an input signal containing the frequency components atfrequency f_(i) or use an internal pilot signal generator set tofrequency f₁; the power of the signal is set to operate the poweramplifier at a nominal operating point;

[0122] (3) set the branch-k CGA gain to some nominal value c′_(k); setall other CGA gains to zero;

[0123] (4) use a partial correlator, with local oscillators set toselect frequency f_(i), to produce the correlation of signal v_(e) withmonitor signal v_(cmk); the result is:

C _(ecmk)(f ₁)=c′_(k) |h _(ck)(f _(i))|² h _(pk) ^(*)(f _(i))·P_(in)(f_(i)), where P _(in)(f _(i))

[0124]  is the input power at frequency f_(i);

[0125] (5) use a partial correlator, with local oscillators set toselect frequency f₁, to produce the correlation of signal monitorv_(cmk) with itself; the result is:

C _(cmk)(f _(i))=|h _(pk)(f _(i))|² ·|h _(ck)(f _(i))|² ·P _(in)(f_(i));

[0126] (6) determine the branch-k observation filter response atfrequency f_(i) as:

h _(pk)(f _(i))c′ _(k) C _(cmk)(f ₁)/(C _(ecmk)(f _(i));

[0127] (7) close the RF switch 1340.

[0128] The observation filter responses for the other filters aredetermined similarly.

[0129] In still another aspect of the present invention, the adjustercircuit 1409 precedes the power amplifier 103, as shown in FIG. 14.Branch filters h_(c0)(v, f) to h_(c,K−1)(v, f) (1430, 1432, 1434) aregeneral nonlinearities with possible frequency dependence, as indicatedby the two arguments v, the input signal, and f, the frequency. Inimplementation, they can take the form of monomial (cubic, quintic,etc.) memoryless nonlinearities. More general nonlinearities such asBessel functions or step functions, or any other convenientnonlinearity, may also be employed. One or more of these branch filtersmay instead have linear characteristics and frequency dependence. Forexample, they may take the form of delays or general linear filters, asin the aspect of the invention described immediately above. In the mostgeneral form, the branch filters depend on both the input signal andfrequency, where such dependencies may be intentional or inadvertent. Inthis model, the amplifier gain is included in the branch filterresponses. The branch filters 1430, 1432, and 1434 respectively precedeCGAs 1431, 1433, and 1435, the outputs of which are summed by combiner1436.

[0130] The filter h_(r)(f) 1410 in the reference branch may also be asimple delay or a more general filter; even if such a filter is notinserted explicitly, h_(r)(f) 1410 represents the response of thebranch. The objective is to determine the responses of the observationfilters h_(p0)(f) to h_(p,K−1)(f) (1420, 1421, and 1422) at selectedfrequencies.

[0131] To determine the response h_(pk)(f_(i)) of the observation filterk at frequency f_(i), the adaptation controller performs the followingactions:

[0132] (1) open the RF switch 1440, thereby disconnecting the filterh_(r)(f_(i)) 1410 from the subtractor 106;

[0133] (2) apply an input signal containing the frequency components atfrequency f_(i) or use an internal pilot signal generator set tofrequency f_(i);

[0134] (3) set all CGA gains other than that for branch k to zero;select the branch-k CGA gain to c′_(k) and the power of the input signalin some convenient combination to cause the power amplifier to operateat a preselected output power that is common to all branches k andfrequencies f_(i) in this calibration procedure; doing so makes theamplifier gain and phase shift the same for all branches and frequenciesduring calibration;

[0135] (4) use a partial correlator, with local oscillators set toselect frequency f_(i), to produce the correlation of signal v_(e) withmonitor signal v_(cmk)(f₁); the result is:

C _(ecmk)(f _(i))=c′ _(k) ·h _(pk) ^(*)(f _(i))·P _(ck)(f _(i)), where P_(ck)(f _(i)) is the

[0136]  power of signal v_(ck) at frequency f_(i);

[0137] (5) use a partial correlator, with local oscillators set toselect frequency f_(i), to produce the correlation of signal monitorv_(cmk)(f_(i)) with itself; the result is:

C _(cmk)(f _(i))=|h _(pk)(f _(i))|² ·P _(ck)(f _(i));

[0138] (6) determine the branch-k observation filter response atfrequency f_(i) as:

h _(pk)(f _(i))=c′ _(k) C _(cmk)(f ₁)/C _(ecmk)(f ₁).

[0139] (7) close the RF switch.

[0140] The observation filter responses for the other filters aredetermined similarly.

[0141] As will be apparent to those skilled in the art in light of theforegoing disclosure, many alterations and modifications are possible inthe practice of this invention without departing from the spirit orscope thereof. For example, the adjuster circuits of an analogpredistorter or a feedforward linearizer can employ both memory andnonlinearity in their branches. A cascade combination of a monomial anda filter within a branch is one way to accomplish this.

[0142] In addition, FIG. 5 illustrates a hybridpredistortion-feedforward circuit. The predistortion adjuster isimplemented with a polynomial adjuster 109 as described above and shownin FIG. 2, and the distortion adjuster is implemented using the FIRadjuster 111 described above and shown in FIG. 4. The signal adjustercircuit 110 is a delay line having a delay selected to match that of thepolynomial adjuster 109 and the power amplifier 103. The delay 112 ofthe delay line in the distortion cancellation circuit 102 is selected tomatch that of the FIR adjuster 111 and error amplifier 108. Thelinearization of the power amplifier may be improved by combining thepredistortion and the feedforward adjuster circuits. Further, ratherthan being a delay line, signal adjuster circuit 110 may be a FIRadjuster.

[0143] Other variations of the preceding linearizer circuits andself-calibration methods are deemed to be within the scope of thepresent invention, which is to be construed solely by the followingclaims.

What is claimed is:
 1. A linearizer for an amplifier, comprising: asignal adjuster having an internal signal; an adaptation controller formonitoring said signal adjuster, the internal signal at an input to saidadaptation controller being deemed a monitor signal, said adaptationcontroller generating a control signal for said signal adjuster byaccounting for a difference between the internal and monitor signals. 2.A linearizer according to claim 1, wherein said adaptation controllerdetermines a response of an observation filter representing thedifference between the internal and monitor signals.
 3. A linearizeraccording to claim 2, wherein said adaptation controller accounts forthe difference between the internal and monitor signals by dividing themonitor signal by the observation filter response.
 4. A linearizeraccording to claim 1, wherein said signal adjuster and amplifier formspart of a signal cancellation circuit of said linearizer.
 5. Alinearizer according to claim 4, wherein said signal adjuster is inseries with the amplifier in the signal cancellation circuit.
 6. Alinearizer according to claim 5, wherein said signal adjuster is inparallel with the amplifier in the signal cancellation circuit.
 7. Alinearizer according to claim 1, wherein said signal adjuster forms partof a distortion cancellation circuit of said linearizer.
 8. A linearizeraccording to claim 1, wherein said signal adjuster comprises an analogpredistorter.
 9. A linearizer according to claim 1, wherein said signaladjuster comprises an FIR filter.
 10. A linearizer according to claim 1,wherein said signal adjuster comprises a filter having a linearcombination of frequency-dependent nonlinearities.
 11. A method forgenerating a control signal for a signal adjuster of an amplifierlinearizer, wherein the signal adjuster has an internal signal,comprising the steps of: monitoring a monitor signal using an adaptationcontroller, the internal signal at an input to the adaptation controllerbeing deemed the monitor signal; generating a control signal for thesignal adjuster by accounting for a difference between the internal andmonitor signals.
 12. A method according to claim 11, further comprisingthe step of determining a response of an observation filter representingthe difference between the internal and monitor signals.
 13. A methodaccording to claim 12, wherein said adaptation controller accounts forthe difference between the internal and monitor signals by dividing themonitor signal by the observation filter response.
 14. A method forself-calibrating a linearizer for an amplifier, the linearizer having asignal adjuster circuit with a plurality of complex gain adjusters, andan adaptation controller, the signal input to each of the complex gainadjusters being output from the signal adjuster circuit to theadaptation controller, said method comprising the steps of: (1) settingthe amplifier to standby so its output is zero; (2) setting a complexgain of a first complex gain adjuster to a nominal value and the othercomplex adjuster gains to zero; (3) applying an input signal to theadjuster circuit, the negative output of the adjuster circuit being anerror signal; (4) bandpass correlating the error signal with the inputsignal to the first complex gain adjuster to generate a firstcorrelation value; (5) bandpass correlating the input signal to thefirst complex gain adjuster with itself to generate a second correlationvalue; (6) computing a first observation filter response, correspondingto the first complex gain adjuster, by multiplying the nominal complexgain value by the second correlation value and by dividing by the firstcorrelation value; (7) repeating steps (2) through (6) to computeobservation filter responses corresponding to the remaining complex gainadjusters, wherein a complex gain is set to a nominal value for eachremaining complex gain adjuster, and the other complex adjuster gainsare set to zero.
 15. A method according to claim 14, further comprisingthe step: (8) adjusting the signal adjuster signals at the input of theadaptation controller by dividing them by the corresponding computedobservation filter responses.
 16. A method for self-calibrating alinearizer for an amplifier, the linearizer having a signal distortionadjuster circuit and a distortion adjuster circuit, the latter circuithaving a plurality of complex gain adjusters, and an adaptationcontroller, the signal input to each of the complex gain adjusters beingoutput from the distortion adjuster circuit to the adaptationcontroller, said method comprising the steps of: (1) setting theamplifier to standby so its output is zero and setting the signaladjuster circuit to produce a non-zero value; (2) setting a complex gainof a first complex gain adjuster to a nominal value and the othercomplex adjuster gains to zero; (3) applying an input signal to theadjuster circuit so as to produce an output signal from the linearizer;(4) bandpass correlating the output signal with the input signal to thefirst complex gain adjuster to generate a first correlation value; (5)bandpass correlating the input signal to the first complex gain adjusterwith itself to generate a second correlation value; (6) computing afirst observation filter response, corresponding to the first complexgain adjuster, by multiplying the nominal complex gain value by thesecond correlation value and by dividing by the first correlation value;(7) repeating steps (2) through (6) to compute observation filterresponses corresponding to the remaining complex gain adjusters of thedistortion adjuster circuit, wherein a complex gain is set to a nominalvalue for each remaining complex gain adjuster, and the other complexadjuster gains are set to zero.
 17. A method according to claim 16,further comprising the step: (8) adjusting the distortion adjustersignals at the input of the adaptation controller by dividing them bythe corresponding computed observation filter responses.
 18. Acontroller for controlling a signal adjuster of a linearizer for anamplifier, the signal adjuster having an internal signal, the controllercomprising: means for computing a difference between the internalsignal, and the internal signal as it exists at an input to saidcontroller (“the monitor signal”); and means for adjusting the monitorsignal based on the computed difference.
 19. A controller according toclaim 18, further comprising means to compute a control signal for thesignal adjuster using the adjusted monitor signal.